1. Field of the Invention
The present invention generally relates to a peak detecting circuit applicable to a spread spectrum technique used for a radio frequency communication system, a distance measurement system or a positioning system and, more particularly, to a correlation peak detecting circuit for detecting a correlation peak in a spread spectrum signal.
2. Description of the Related Art
FIG. 1 is a block diagram of a digital matched filter 200 used in a conventional spread spectrum circuit. The digital matched filter 200 shown in FIG. 1 comprises a shift register 201, a plurality of multipliers 202 each of which multiplies a signal input from the shift register 201 by a tap coefficient and a summing circuit 203 which sums output signals of the multipliers 202. Normally, the tap coefficient is either +1 or xe2x88x921 when a correlation of a pseudo noise (PN) signal is obtained.
As appreciated from FIG. 1, since the digital matched filter 200 is constituted by a sequence circuit including the shift register 201 and other parts, a time discrete signal which may be produced by sampling by an A/D converter and the like must be input to the digital matched filter 200.
When the PN signal is subjected to a discrete signal producing process such as a sampling process, the autocorrelation characteristic of the processed PN signal differs from the original characteristic. FIGS. 2A, 2B and 2C show examples of correlation characteristics represented by a correlation output with respect to a phase shift xcex94xcfx84 from a reference phase. FIG. 2A shows an example of an original correlation characteristic; FIG. 2B shows a correlation characteristic when a discrete PN signal is input; FIG. 2C shows a correlation characteristic when a discrete PN signal produced by sampling under a bandwidth restriction is input. It should be noted that, in FIG. 2B, two samples are taken per one chip timing. For the sake of convenience, two samples are taken per one chip timing for all cases described below.
FIGS. 3A to 3E show examples of outputs of a matched filter when a discrete PN signal is input to the matched filter.
FIG. 3A shows an example of output of an analog matched filter. As shown in FIG. 3A, the analog matched filter outputs a signal pulse for each single cycle of the input PN signal.
FIG. 3B shows an example of output of a digital matched filter. In the example of FIG. 3B, the envelope of the signals output from the digital matched filter is the same as that of the analog matched filter shown in FIG. 3A. However, the output itself is discrete. This is because a shift register of the digital matched filter also performs a discrete shifting process in synchronization with a sampling clock of the A/D converter.
Accordingly, a largest peak signal and two peak signals having a level which is one half of the largest peak signal must always be obtained from the correlation output using the digital matched filter for each cycle as shown in FIG. 3B irrespective of a sampling timing. However, in practice, the signal to be input to the digital matched filter is influenced by a bandwidth restriction.
FIG. 4A shows an example of the PN signal transmitted by a sender. FIG. 4B shows a correlation characteristic when the PN signal shown in FIG. 4A is input to the digital matched filter.
As shown in FIG. 4B, the PN signal input to the digital matched filter is blunted or dulled due to a bandwidth restriction according to a legal regulation or a system performance. The blunted or dulled signal is subjected to a sampling process, and a waveform as shown in FIG. 4C or 4D is obtained and input to the digital matched filter.
Accordingly, the correlation characteristic of the sampled PN signal becomes different from that of the input signal shown in FIG. 4B. Thus, in a case of the digital matched filter, the output of the digital matched filter becomes as shown in FIG. 3C. Naturally, the output characteristic of the digital matched filter can be a characteristic as shown in either FIG. 3D or FIG. 3D which has an envelop equivalent to the characteristic shown in FIG. 3C.
As appreciated from the above-mentioned example, when the PN signal subjected to the bandwidth restriction is sampled and is input to the digital matched filter, the correlation output varies according to the sample timing and the output signal pulses do not always represent a value of the correlation peak. Additionally, there is a problem in that a time when a largest peak pulse from among the signal pulses appears is varied.
Accordingly, in a communication system merely using a digital matched filter, it is required to set a threshold value for detecting the correlation peak to match a lowest value of the correlation output. Additionally, a time when a correlation peak appears cannot be estimated. This results in deterioration in the transmission characteristic.
In the spread spectrum communication, an information signal is transmitted by being multiplied by a PN signal having a sufficiently high speed. On a receiver side, the information signal spread by the PN signal is processed by a matched filter or a sliding correlator so as to enable a demodulation process.
Particularly, in a case in which the reception signal is input to the matched filter, a code system the same as the PN signal used for spreading the information signal on the sender side is set to a set of coefficients used by the matched filter. Thereby, when the spread information signal is input to the matched filter, the matched filter outputs peak signals having a sharp peak as shown in FIG. 5A. The receiver side detects a time when the peak signal appears so as to detect a phase of the received signal.
However, FIG. 5A shows an ideal case, and, in practice, the received signal is influenced by a bandwidth restriction and the waveform of the received signal is blunted or dulled. Accordingly, the correlation characteristic becomes as shown in FIG. 5B.
In order to constitute a matched filter, an analog system using a SAW filter and the like or a digital system can be used. The digital system has an advantage over the analog system with respect to cost and size since the digital system can be achieved by an integrated circuit.
FIG. 6 shows a structure of a conventional digital matched filter (DMF). The digital matched filter shown in FIG. 6 comprises a plurality of delay elements 211, a plurality of multipliers 212 and an adder 213 which sums outputs of the amplifiers 212. Each of the delay elements 211 delays an inputting timing corresponding to a single cycle. Each of the multipliers 212 multiplies an output of the corresponding delay element by a coefficient hi (i=1 to m). The coefficient hi takes either a value of +1 or xe2x88x921.
Since the DMF is constituted by a digital circuit, the signal input thereto is a discrete signal which is obtained by sampling the received signal at every predetermined time. Additionally, the received signal is quantized in response to a dynamic range of the input signal. Hereinafter, an i-th sampled signal with respect to a reference time is represented by Xi.
The signal input to the DMF is delayed by a multi-bit shift register, and the following signals are output from the respective shift registers, where m is a number of shift registers.
{Xixe2x88x921, Xixe2x88x922, Xixe2x88x923, . . . , Xixe2x88x92m}
The output of each of the shift registers is multiplied by the respective coefficient, and summed by the adder 203. Accordingly, the output signal yi of the DMF is represented as follows.       Y    i    =            ∑              j        =        1            m        ⁢                  h        i            ⁢              x                  i          -          j                    
Accordingly, the output of the DMF is also the discrete signal Yi. The output characteristic of the output Yi is a train of discrete signals as shown in FIG. 5C or FIG. 5D. That is, the train of signals shown in FIG. 5C or FIG. 5D is obtained by sampling the correlation characteristic shown in FIG. 5B. In the conventional technique, it is determined that the peak of the correlation characteristic appears at a time when a largest value of the signals appears in the characteristic shown in FIG. 5C or FIG. 5D.
When the conventional DMF is used, there are following problems.
1) When a dynamic range of the input signal is large, a large number of quantization bits of each shift register are required so as to represent the dynamic range. Additionally, a large number of bits are required for an arithmetic circuit in the DMF so as to handle pulse signals. Accordingly, there is a problem in that a large and complex circuit is required and an operation speed is reduced.
2) There is a possibility that an offset of the input signal fluctuates due to a temperature change and a change with respect to elapsed time.
3) When a correlation characteristic shown in FIG. 5B is represented by discrete signals, the output signal differs as shown in FIGS. 5C and 5D according to sample timing. This prevents an accurate detection of a peak.
Japanese Laid-Open Patent Application No. 9-501032 discloses a method for receiving and decoding communication signals in a CDMA receiver. In this method, the CDMA receiver is provided with a function of a digital matched filter equivalent so as to suppress influences of the correlation output to signals of other channels. However, in this method, the tap coefficients of the digital matched filter must be represented by a plurality of numbers each of which is represented by a plurality of bits. Accordingly, each of the multipliers in the digital matched filter must have a function of (multi-value input signal)xc3x97(multi-value tap coefficient). Thus, there is a problem in that the size of the circuit is large.
It is a general object of the present invention to provide an improved and useful peak detecting circuit and method in which the above-mentioned problems are eliminated.
A more specific object of the present invention is to provide a peak detecting circuit and method capable of accurately detecting a peak of a time discrete signal without increasing a circuit scale.
In order to achieve the above-mentioned object, there is provided according to the present invention a peak detecting circuit for detecting a peak of a waveform of a time discrete signal, the peak detecting circuit calculating an approximate function which approximates the waveform of the time discrete signal so as to detect a peak of the approximate function so that the peak of the approximate function is estimated as the peak of the waveform of the time discrete signal.
According to the present invention, the time discrete signal having a sharp peak is represented by the approximate function. The peak of the time discrete signal is detected as an extremum of the approximate function. Sine the approximate function can be represented by a quadratic polynominal, a peak value or a time when the peak value appears can be estimated by a simple calculation based on coefficients of the quadratic polynominal. Thus, the peak detecting circuit according to the present invention can be achieved by a simple circuit structure.
In one embodiment of the present invention, the time discrete signal is output from a digital matched filter to which a signal obtained by sampling a pseudo noise signal is input. The pseudo noise signal is subjected to a bandwidth restriction during transmission through a communication medium. The digital matched filter outputs the time discrete signal which represents an autocorrelation characteristic of the pseudo noise signal. According to the present invention, the autocorrelation characteristic is approximated by approximate function and a peak of the approximate function is estimated as a peak of the autocorrelation characteristic. Thus, a peak of the original pseudo noise signal can be obtained without being influenced by sample timing applied to the pseudo noise signal.
According to the present invention, a time when a correlation peak appears can be estimated by parameters of the approximate function. Additionally, a phase difference between sample timing and the pseudo noise signal can be detected based on the estimated time of appearance of the correlation peak. Thus, the sample timing can be matched to the time of appearance of the correlation peak. Additionally, since the time when the next correlation peak appears can be estimated, an output of the digital matched filter can be permitted only when the next correlation peak appears. That is, a so called window function can be applied to the correlation output. This eliminates influence of noise appearing during an interval of the correlation peaks.
In practice, when the phase difference between the output of the digital matched filter and the sampled pseudo noise signal input to the digital matched filter is detected, a difference less than one cycle of a system clock is rarely needed. Thus, the phase difference can be calculated on a unit time basis corresponding to the one clock cycle. Thus, the phase difference can be represented by an integer.
In one embodiment of the present invention, the approximate function is represented by a quadratic polynominal. This allows the peak detecting circuit to be a simple, compact hardware structure which results in a high-speed operation and a good transmission characteristic.
When a digital matched filter is used to construct a communication system, detection of the correlation peak value itself is not required in many cases. In such a case, presence of the correlation peak or the plus or minus sign of the correlation peak value can be determined by a simple calculation using parameters of the approximate function without calculating the correlation peak value itself. This eliminates erroneous detection of the correlation peak due to influence of fluctuation in an offset of the correlation output or sample timing
In the present invention, the correlation peak is detected according to the parameters of the approximate function by using a threshold value. The threshold value may be set based on previously obtained parameters of the quadratic function. Accordingly, the threshold value can be adaptively set with respect to fluctuation in the correlation peak due to fluctuation in a transmission path characteristic and the like.
In one embodiment of the present invention, a differential signal may be input to the digital filter so that a dynamic range of the signal input to the digital matched filter is decreased. When such a differential signal is input to the digital matched filter, a pulse-like signal does not appear in the signal input to the digital matched filter. Thereby, a number of bits used by calculations in the digital matched filter is reduced. This reduces a size and power consumption of the digital matched filter. Additionally, operation logic can be simplified which results in an increase in an operation speed.
Additionally, by using the differential signal, an offset of the signal input to the digital matched filter can be cancelled.
When the digital matched filter is used, there may be a case in which information with respect to a sign of the correlation peak value is required. According to the present invention, a sign of the correlation peak value can be determined based on a curvature of the approximate function. The curvature can be represented by a result of a simple calculation using parameters of the approximate function. The result of determination of the sign of the correlation peak value may be used for demodulation of data output from the digital matched filter.
If the digital matched filter is used in a code division multiple access system (CDMA), the output of the digital matched filter is influenced by a cross-correlation between other communication channels. This may cause an erroneous detection of a correlation peak. The present invention eliminates such an erroneous detection by using a curvature of the approximate function as a condition for determining presence of the correlation peak.
Additionally, there is provided according to another aspect of the present invention a peak detecting method for detecting a peak of a waveform of a time discrete signal, the peak detecting method comprising the steps of:
calculating an approximate function which approximates the waveform of the time discrete signal; and
calculating a peak time when the peak of the waveform of said time discrete signal appears.
According to the above-mentioned invention, the time discrete signal having a sharp peak is represented by the approximate function. The peak of the time discrete signal is detected as an extremum of the approximate function. Sine the approximate function can be represented by a quadratic polynominal, a peak value or a time when the peak value appears can be estimated by a simple calculation based on coefficients of the quadratic polynominal. Thus, the peak detecting method according to the present invention can be achieved by a simple circuit structure.
The peak detecting method according to the present invention may further comprise the steps of:
inputting a signal obtained by sampling a pseudo noise signal transmitted via a communication medium to a digital matched filter; and
obtaining the time discrete signal from the digital matched filter.
Additionally, the peak detecting method according to the present invention may further comprise the steps of:
calculating parameters of the approximate function;
calculating an extremum of the approximate function; and
determining the calculated extremum to be a peak value of the time discrete signal.